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power management 30v, 300ma output micropower step-down switching regulator typical application circuit september 24, 20 2 applications portable equipment notebook computers distributed supplies backup power supplies ? ? ? ? 5v output step-down converter figure 1. SC4530 features input voltage range: 3v to 30v low quiescent current: drawing 9 m a from v in when stepping down from 2v to 3.3v at no load high efficiency from 2v input to 5v output > 80% at 650 m a > 85% at 0ma - 300ma up to 300ma continuous dc output current integrated power switch and schottky diodes low output ripple < m a shutdown current hysteretic current-mode control cycle-by-cycle current limiting alternating between micropower idling and switch - ing states at light loads to conserve power output short-circuit protection solution footprint as small as 50mm 2 low-profile 3mm x 2mm mlpd 8-lead package ? ? ? ? ? ? ? ? ? ? ? ? ? description the SC4530 is a micropower hysteretic current-mode step-down switching regulator capable of providing up to 300ma of output current from 3v to 30v input voltage range. it is designed to provide very high standby efficiency while simplifying design. at light loads, the SC4530 switches only as needed to main - tain regulation, while idling most of the time, to improve efficiency. typical quiescent currents from v in and bias are 7 m a and 26 m a r espectively. the control scheme pro - duces less than 0mv of fb voltage ripple at light loads. the SC4530 automatically switches to continuous-conduction mode at heavy loads. the SC4530 has integrated power devices and on-chip control circuitry, simplifying design and enabling a solu - tion footprint as small as 50mm 2 . only an inductor and a few passive components are needed to complete a dc-dc regulator. the inductor current hysteretic control of SC4530 makes it inherently short-circuit robust. the wide input voltage range enables the device to operate from a variety of input sources, including single- or multi-cell batteries, system rails and wall transformers. efficiency vs load current 40 50 60 70 80 90 0.1 1.0 10.0 100.0 1000.0 load current (ma) efficiency (%) v in = 12v v in = 24v v out = 5v l1: coilcraft lps4018 c2: murata grm32er71h475k c1: murata grm31cr60j226k SC4530 sw bst in bias out 5v/0.3a r1 619k v in 200k r2 l1 33 h en c3 0.22 f c2 4.7 f gnd fb c1 22 f c4 10pf off on c4=22pf c5=68pf verified 1/24/2011 c4=10pf c5=33pf verified 1/12/2012
2 pin confguration marking information ordering information notes: () available in tape and reel only. a reel contains 3,000 devices. (2) available in lead-free package only. device is weee and rohs compliant. ja = 80 c/w mlpd: 3mm x 2mm 8 lead gnd bias in nc en bst sw fb 9 8 7 6 5 1 2 3 4 SC4530 device package SC4530wltrt () (2) mlpd-w-8 3x2 SC4530evb evaluation board xxxx - lot number 4530 xxxx
3 exceeding the above specifcations may result in permanent damage to the device or the device may malfunction. operation outside of the parameters specifed in the electrical characteristics section is not recommended. notes: () unless noted otherwise, all voltage values in this section are with respect to ground. (2) tested according to jedec standard jesd22-a 4-b. (3) calculated from package in still air, mounted to 3 x 4.5, 4-layer fr4 pcb with thermal vias under the exposed pad, per jesd5 standards. electrical characteristics unless otherwise noted, t a = 25 c for typical values, -40 c < t a = t j < 25 c. v in = v en = 0v, v bst = 5v, v bias = 3v. absolute maximum ratings (1) in -0.3v to 32v sw -0.6v to v in bst 42v bst above sw 30v fb -0.3v to .9v bias -0.3v to v in en -0.3v to v in esd protection level (2) 2kv recommended operating conditions thermal resistance, junction to ambient (3) 80c/w maximum junction temperature+ 50 c storage temperature range -65c to + 50c peak ir reflow temperature ( 0s to 30s) +260c thermal information SC4530 parameter name conditions min typ max units v in operating range 3 30 v v in quiescent supply current v en = 0.2v 0. 0.5 a not switching 7 a not switching , v bias = 0 34 50 a bias quiescent supply current v en = 0.2v 0. 0.5 a not switching 26 40 a not switching , v bias = 0 0. a en pin input high voltage 2 v en pin input low voltage 0.2 v en pin current v en = 2.5v 2.5 a feedback voltage v fb falling .22 .232 .245 v feedback voltage line regulation v in = 3v to 30v 0.0 0.02 %/v fb pin bias current v fb = .25v 20 60 na minimum switch of-time t off(min) 530 ns maximum switch duty cycle d max 90 96 % junction temperature range -40c to + 25c v in 3v to 30v output dc current up to 300ma
4 electrical characteristics (continued) pin descriptions pin # pin name pin function fb inverting input of the error amplifer. the fb pin is tied to a resistive divider between the output and ground. the voltage divider sets the output voltage. 2 bias anode of the internal bootstrap diode. bias also powers the internal control circuit if v bias > 2.3v. tie to the output of the dc-dc converter if v out > 2.5v. tie bias to in if v out is set below 2.5v. 3 bst power transistor driver supply. connect an external bootstrap capacitor from the sw pin to this pin to generate a drive voltage higher than v in to fully saturate the internal power transistor. 4 sw the power transistor emitter and the cathode of the freewheeling diode. the sw pin is connected to an inductor and a bootstrap capacitor. 5 gnd connect this pin to the pc board power ground plane. 6 in power supply to the SC4530. it must be closely bypassed to the ground pin. 7 nc no connection. 8 en the enable pin for the SC4530. driving this pin below 0.2v completely shuts of the SC4530. applying more than 2v to this pin enables the SC4530. if not driven from a control circuit, tie this pin to in. 9 exposed pad the exposed pad at the bottom of the package serves as a thermal contact to the circuit board. it is to be soldered to the ground plane of the pc board. SC4530 parameter name conditions min typ max units switch current limit i lim v fb = 0 0.39 0.50 0.66 a inductor current hysteresis () v fb = 0 65 ma switch saturation voltage i sw = -0.3a 200 300 mv switch leakage current v sw = 0 2 a switch minimum bootstrap voltage i sw = -0.3a .7 2.2 v bst pin current i sw = -0.3a 7. 2 ma freewheeling diode forward voltage v d i sw = -0.3a 700 mv freewheeling diode reverse leakage v sw = 0v 5 a bootstrap diode forward voltage i bst = 40ma 700 mv bootstrap diode reverse leakage v sw = 0v, v bias = 0 a notes: ( ) the inductor current hysteresis is the difference between the switch current limit and the freewheeling diode valley current. unless otherwise noted, t a = 25 c for typical values, -40 c < t a = t j < 25 c. v in = v en = 0v, v bst = 5v, v bias = 3v.
5 block diagram figure 2. SC4530 block diagram SC4530 in 6 q1 ipk + - v os bandgap reference bst 3 sw 4 fb 1 cmp + - gnd 5 ea + - oc en 8 ivly + - uc 1.232v r s1 bias 2 d1 r q s q d2 530ns min. t off run / idle 530ns r r s2 i cntl r dz v hys } parasitics
6 peak and valley current dc thresholds vs temperature (1) 0.30 0.35 0.40 0.45 0.50 0.55 0.60 -50 -25 0 25 50 75 100 125 temperature ( o c) current (a) v bst = 15v v in = 10v valley peak v fb = 0 SC4530 switch saturation voltage vs switch current 0 100 200 300 400 0.0 0.1 0.2 0.3 0.4 0.5 0.6 switch current (a) saturation voltage (mv) v in = 10v v bst = 15v 125 o c -55 o c 25 o c bst pin current vs switch current 0 5 10 15 0.0 0.1 0.2 0.3 0.4 0.5 0.6 switch current (a) bootstrap current (ma) v in = 10v v bst = 15v 25 o c -55 o c 125 o c typical characteristics minimum off time vs temperature 400 450 500 550 600 650 700 -50 -25 0 25 50 75 100 125 temperature ( o c) off time (ns) v bst = 15v v in = 10v maximum on time vs temperature 10 11 12 13 14 15 -50 -25 0 25 50 75 100 125 temperature ( o c) on time ( p s) v bst = 15v v in = 10v minimum bootstrap voltage vs temperature 1.2 1.4 1.6 1.8 2.0 2.2 -50 -25 0 25 50 75 100 125 temperature ( o c) v bst - v sw (v) i sw = -0.39a notes: ( ) circuit propagation delays and the error amplifer output voltage ripples may cause the actual inductor valley current to differ from its dc value. efficiency vs load current v out = 3.3v 30 40 50 60 70 80 90 0.1 1.0 10.0 100.0 1000.0 load current (ma) efficiency (%) v in = 5v v in = 24v v in = 12v coilcraft lps4018-333ml efficiency vs load current v out = 2.5v 30 40 50 60 70 80 90 0.1 1.0 10.0 100.0 1000.0 load current (ma) efficiency (%) v in = 5v v in = 24v v in = 12v coilcraft lps4018-333ml feedback voltage vs temperature 1.20 1.21 1.22 1.23 1.24 -50 -25 0 25 50 75 100 125 temperature( o c) feedback voltage (v) v in = 10v
7 SC4530 quiescent currents vs bias voltage 0 10 20 30 40 50 0.0 1.0 2.0 3.0 4.0 bias voltage (v) current ( p a) 125 o c v in = 10v -40 o c -40 o c i in i bias quiescent currents vs v in 0 5 10 15 20 25 30 0 5 10 15 20 25 30 v in (v) current ( p a) i bias 25 o c i in v bias = 3v bias quiescent current vs temperature 0 10 20 30 40 -50 -25 0 25 50 75 100 125 temperature ( o c) current ( p a) v bias = 3v v in = 10v v in quiescent current vs temperature 0 10 20 30 40 -50 -25 0 25 50 75 100 125 temperature ( o c) current ( p a) v bias = 1v v in = 10v v bias = 3v freewheeling diode forward characteristics 1 10 100 1000 0.2 0.4 0.6 0.8 1.0 voltage (v) current (ma) -40 o c 25 o c 125 o c freewheeling diode reverse leakage current 0 5 10 15 0 5 10 15 20 25 30 35 reverse voltage (v) reverse current ( p a) 125 o c bootstrap diode forward characteristics 0.1 1.0 10.0 100.0 0.2 0.4 0.6 0.8 1.0 voltage (v) current (ma) -40 o c 25 o c 125 o c typical characteristics (continued) en pin current vs v en 0.0 0.5 1.0 1.5 2.0 0 5 10 15 20 25 30 v en (v) current ( p a) -40 o c 25 o c 125 o c bootstrap diode reverse leakage current 0 1 2 3 4 5 0 5 10 15 20 25 30 35 40 v bst - v bias (v) reverse current ( p a) 125 o c
8 SC4530 general description and operation the SC4530 is a micropower, hysteretic current-mode step-down switching regulator. as shown in the block diagram in figure 2, the converter is controlled by an error amplifer ea and two current-sensing comparators ipk and ivly. ipk and ivly monitor the switch (q ) collector current and the freewheeling diode (d ) current respectively. the ea amplifes the diferential voltage between the fb and the bandgap reference, and produces a current, i cntl , proportional to its output voltage. i cntl , in turn, adjusts the switching thresholds of both the peak and valley current comparators. the ea output voltage is high at heavy loads, as is the peak inductor current. the zener diode d z clamps the amplifer output and sets the switch peak current limit. when the switch q is turned on, the current through q ramps up until it reaches the peak threshold set by i cntl . the output of the ipk comparator, oc, goes high. this resets the latch and turns of the switch. with q of, the inductor current ramps down through the freewheeling diode d . when d current ramps below the valley threshold established by i cntl , the output of the ivly comparator, uc, goes high. if q has been turned of for more than 530ns, then the latch will be set and q will again turn on, starting a new cycle. the inductor ripple current in continuous-conduction mode is independent of i cntl and is primarily determined by v os and v hys . continuous mode switching frequency, therefore, depends on v in , v out , the inductance l and the propagation delay times of the current comparators. if the regulator output is shorted to ground, then the amplifer output will rise to d z clamp voltage. q turns of as the inductor current reaches the peak current limit. with the output shorted to ground, the inductor current ramps down at a slower rate through d . q turns on again when the inductor current crosses the valley threshold. therefore, short-circuiting the output merely lowers the converter switching frequency. the inductor current remains bounded by the peak switch current limit. the run/idle comparator, cmp, monitors the output of the error amplifer. if the ea output falls below the run/ idle threshold, then q and all control circuits except the reference and ea will be shut of. the output capacitor will then supply the load, causing the output voltage to fall. when the ea output rises above the run/idle threshold, the control circuit wakes up and the part starts to switch, delivering power to the output. the ofset voltage v os at the input of the ipk comparator ensures that any current pulse delivered to the output has some minimum amplitude. at very light loads, even a single minimum charge packet delivered to the output will cause the fb voltage to rise above the reference voltage. this causes the ea output voltage to fall and the part to idle. the part resumes switching when the output current discharges the fb voltage below the reference. at light loads, the part switches only as needed to keep the output in regulation. by reducing the supply current drawn when idling, high efciency is maintained at light loads. at heavier loads, it may take a number of consecutive minimum pulses to bring the fb above the reference voltage. the part enters continuous conduction mode when the amplifer output never falls below the run/idle threshold. driving the base of the power transistor above the input power supply rail minimizes the power transistor turn- on voltage and maximizes efciency. a bootstrap circuit [formed by an internal bootstrap diode d 2 (figure 2) and an external capacitor connected between bst and sw] generates a voltage higher than v in at the bst pin. the bootstrapped voltage becomes the supply voltage of the power transistor driver. the internal control circuit takes its power from either the input or from the bias pin if v bias > 2.3v. for applications with output voltage higher than 2.5v, the bias pin should be tied to the regulator output to maximize efciency.
9 applications information SC4530 setting the output voltage the SC4530 output voltage is programmed using a resistive divider (figure 3) with its center tap tied to the fb pin. for a given r 2 , r can be determined: ? 1 ? ? 1 232 . 1 v r r out 2 1 ( ) the percentage error due to the input bias current of the error amplifer is: v 232 . 1 ) r r ( 100 na 20 v v 2 1 out out ? ? ? ' (2) example: determine the output voltage error caused by the amplifer input bias current in a 5v output converter. assuming r 2 = 200k w and using equations ( ) and (2), : | ? 1 ? ? : k 619 1 232 . 1 5 k 200 r 1 % 25 . 0 v 232 . 1 ) k 619 k 200 ( 100 na 20 v v out out ? ? ? ' using large r and r 2 helps in maintaining light-load efficiency, since the current drawn by the feedback resistive divider is not delivered to the converter output. the simple calculation above shows that relatively large r and r 2 can be used without introducing more error than that resulting from the tolerance of the standard % resistors. maximum duty cycle limitation the SC4530 is a non-synchronous, step-down switching regulator. its duty cycle in continuous-conduction mode (ccm) primarily depends on the input and output voltages: cesat d in d out v v v v v d (3) where v cesat = 0.25v is the switch saturation voltage and v d = 0.6v is the forward voltage drop of the freewheeling diode. whenever the power switch is turned of, it is kept of for at least 530ns. moreover, the control circuit prevents the power transistor from turning on for more than 3.5 m s. the inductor current pulls the sw node low as the power switch turns of, allowing the inductor current to charge the bootstrap capacitor. the maximum on-time ensures that the bootstrap capacitor gets replenished after a long switch-on interval. the minimum of-time, together with the maximum on-time, put an upper limit on the achievable duty cycle ( 0.96). from equation (3), the minimum v in to avoid dropout is: d cesat d out ) min ( in v v 96 . 0 v v v (4) if v in falls below this minimum, then the regulator will not be able to attain its set output voltage regardless load. using equation (4), the input supply voltage must be at least 5.5v in order to generate a 5v output. inductor selection the SC4530 uses a hysteretic current-mode control topology. the peak-to-peak inductor ripple current, ?i l , is theoretically constant. however, propagation delays of the current comparators (ipk and ivly in figure 2), as well as the error amplifer (ea) output ripples, will cause the actual inductor ripple current to vary depending on the input voltage and the duty cycle. the inductor should be chosen so that the valley current comparator, not the minimum of-time, determines the switch turn-on instant. to simplify inductance calculation, we will assume that ?i l is constant and equal to 50ma. furthermore, we will use .5 times the typical t off(min) , to allow for tolerance and temperature variation. l ) min ( off d out min i t 5 . 1 v v l ' ? (5) figure 3. r 1 and r 2 set the output voltage r2 r1 20na fb SC4530 v out r2 r1 20na fb SC4530 v out
0 applications information (continued) SC4530 for a given v in and inductance l, the continuous- conduction switching frequency is: (6) the minimum inductance is frst found using equation (5). next the switching frequencies are estimated at v in extremes using equation (6). the inductance is then adjusted for achieving desired switching frequency. the resulting switch on-time at the maximum v in must exceed the minimum controllable switch on-time, which can be as high as 80ns. this prevents the inductor current from running away when the output is shorted to ground. example: select the inductor for a 3.3v output regulator, with input voltage ranging from 0v to 26v. the desired switching frequency is about 600khz. the minimum inductance is found using equation (5): duty cycles and switching frequencies at the input voltage extremes can be found using equations (3) and (6) respectively. the results for 22 m h and 33 m h are tabulated (table ). table 1. estimated switching frequencies for 3.3v output input voltage v in (v) duty cycle d (%) switching frequency f (khz) l = 22h l = 33h 0 37.7 740 490 26 4.8 000 670 the 33 m h inductance will be chosen, as it gives the desired switching frequency range. the resulting switch on-time is checked against the minimum controllable switch on-time. the switch on- time can be calculated using equation (7) below: (7) with l = 33h, the switch on-time at 0v and 26v v in are 770ns and 220ns respectively, above the 80ns minimum controllable on-time. table 2 lists some recommended inductor values for various output voltages. table 2. recommended inductor values output voltage v out (v) inductor value ( h) v in = 6 v v in = 30 v .8 22 47 2.5 22 33 3.3 22 33 5.0 33 33 2 68 68 8 - 00 the saturation current of the inductor should be at least 20%~30% higher than the peak inductor current. low- cost inductors with powder iron cores are not suitable for high-frequency switching due to their high core losses. inductors with ferrite cores are recommended. input capacitor selection a step-down regulator draws pulse current from the input power supply. a capacitor placed between the supply and the converter flters the ac current and keeps the current drawn from the supply to a dc constant. the input capacitance should be high enough to flter the pulse input current. its equivalent series resistance (esr) should be low so that power dissipated in the capacitor does not result in signifcant temperature rise and degrade reliability. multi-layer ceramic capacitors, which have very low esr (a few m w ) and can easily handle high rms ripple current, are the ideal choice. a single 4.7f (x5r or x7r) ceramic capacitor should be enough for most applications. using a larger capacitor (for example, 0f) will reduce sw node jitters if the minimum input voltage is less than 0.7v above the output voltage. for applications with high input voltage, a small ( f ~ 2.2f) ceramic capacitor can be placed in parallel with a low esr electrolytic capacitor to satisfy both the esr and bulk capacitance requirements. l cesat d in cesat out in d out l d out i ) v v v ( l ) v v v )( v v ( i l ) d 1 )( v v ( f ' ' ? h 22 15 . 0 5 . 1 53 . 0 ) 6 . 0 3 . 3 ( l min p | u u out cesat in l on v v v i l t ' ?
SC4530 applications information (continued) output capacitor selection the output ripple voltage ?v out of a step-down regulator in continuous conduction can be expressed as: ? ? 1 ? ' ' out off on l out c 8 t t esr i v (8) where c out is the output capacitance. the frst term in equation (8) results from the equivalent series resistance (esr) of the output capacitor while the rest is due to the charging and discharging of c out by the inductor ripple current. substituting ?i l = 50ma, t on + t off = 2s, c out = 22f and esr = 3m in equation (8), we get: mv 2 . 2 7 . 1 45 . 0 ) m 4 . 11 m 3 ( a 15 . 0 v out : : ? ' depending on the switching period and the type of the capacitor used, the output voltage ripple resulting from charging/discharging of c out may be higher than the ripple due to the esr. the example above also shows that the output voltage ripple in continuous mode is very low. the SC4530 relies on fast amplifer response to reduce the output voltage overshoot during power-up. neither the error amplifer output nor the reference is ramped during start-up. the zener diode d z (refer to page 5) clamps the amplifer output, while the regulator output voltage ramps up. as a result, the switch q is turned of every cycle at the switch current limit, i lim (typically 0.5a). the regulator thus delivers about 0.5a to its output until v out rises to its set value. if the load is light, then the amplifer output voltage will fall below the run/idle threshold following regulation. this causes the regulator to idle. however the energy previously stored in the inductor still fows to the output, causing the output voltage to rise above its regulation level. the minimum output capacitance required to keep the overshoot to less than % of the nominal output voltage is: d out out 2 lim out v v v i l 50 c ! (9) the minimum output capacitance for various output voltages can be estimated from equation (9) using the inductances given in table 2. the results are shown in table 3. smaller output capacitors may also be used if higher output voltage overshoot is acceptable. table 3. calculated minimum output capacitance for 1% v out overshoot during start-up v out (v) minimum c out (f) v in = 6v v in = 30v .8 64 96 2.5 36 53 3.3 22 33 5.0 5 5 2 5.6 5.6 8 - 3.7 ceramic capacitors are the best choice for most applications. sanyo tpe series polymer capacitors in b- case, which ofer large capacitors (> 00f) with slightly higher esr, are also good alternatives. ripple current in the output capacitor is not a concern because the inductor current of a step-down converter directly feeds c out , resulting in very low ripple current. avoid using z5u or y5v ceramic capacitors because these types of capacitors have high temperature and high voltage coefcients. bootstrapping the power transistor to reduce the switch on-state voltage and maximize efficiency, the base of the power transistor should be driven from a power supply higher in voltage than v in . the required driver supply voltage (at least 2v higher than the sw) is generated with a bootstrap capacitor c 3 connected between the bst and the sw nodes (figure ) and the bootstrap diode d 2 (figure 2). the d 2 anode is connected to the bias pin. during startup, the power transistor in the SC4530 is first switched on so the current flows through to the inductor. when the transistor is switched off, the inductor current pulls the sw voltage low, allowing c 3 to be charged through the internal bootstrap diode d 2 . when the power switch is turned on again, the sw voltage goes high. this brings the bst voltage to v sw + v c3 , thus back-biasing d 2 .
2 SC4530 applications information (continued) the c 3 voltage increases with each subsequent switching cycle, as does the bootstrapped voltage at the bst pin. after a number of switching cycles, c 3 will be fully charged to a voltage approximately equal to that applied to the anode of d 2 . the minimum bst to sw voltage required to fully saturate the power transistor is shown in the typical characteristics (pages 6-7). this difference voltage must be at least .72v at room temperature. this is also specified in the electrical characteristics (pages 3-4) as the minimum bootstrap voltage. the minimum required v c3 increases as temperature decreases. the bootstrap circuit reaches equilibrium when the base charge drawn from c 3 during transistor on-time is equal to the charge replenished during the off interval. figure 4 summarizes various ways of bootstrapping the SC4530. in figure 4(a) the bias pin is connected to the converter output. the bootstrap charge is obtained from the output of the step-down converter. the input- referred charge is reduced by the step-down ratio. this is the most efficient configuration and it also results in the least voltage stress at the bst pin. the maximum bst pin voltage is about v in + v out . if the output voltage is between 2.5v and 3v, then a 0.33-0.47 m f bootstrap capacitor may be needed to reduce droop. in most other cases, a 0.22 m f ceramic capacitor is adequate. figure 4(b) shows the SC4530 can also be bootstrapped from the input. this way it is not as efficient as the configuration shown in figure 4(a). however this may be only option if the output voltage is less than 2.5v and there is no other supply with voltage higher than 2.5v. voltage stress at the bst pin can be somewhat higher than 2v in . the bst pin voltage should not exceed its absolute maximum rating of 42v. figure 4(c) shows how to bootstrap the SC4530 from an independent power supply v s with its voltage > 2.5v. to demonstrate the effect of an under-sized bootstrap capacitor, c 3 (figure , page ) is deliberately reduced to 0nf. the bias pin is tied to an external power supply similar to figure 4(c). by adjusting the external supply voltage (d) (e) (c) v s > 2.5v SC4530 bst gnd in sw bias 0.1 f v out v in c 3 max v bst ? v in + v s SC4530 bst gnd in sw d z - v z + bias v out v in c 3 v bst ? 2v in - v z v in ? v z > 2.5v (a) SC4530 bst gnd in sw c 3 v in bias max v bst ? v in + v out v out > 2.5v (b) SC4530 bst gnd in sw bias c 3 v in max v bst ? 2v in v out < 2.5v v out > v z + 2.5v SC4530 bst gnd in sw c 3 v in bias d z - v z + v bst ? v in + v out - v z 0.1 f figure 4. methods of bootstrapping the SC4530
3 SC4530 v s , the bootstrap voltage can be varied. figure 5(a) shows the switching waveforms of a correctly bootstrapped 0v to 5v regulator with v s = 2.5v. all three traces share the same ground level. when the power transistor is turned on, v sw should come within a few hundred millivolts of v in applications information (continued) and v bst should have at least 2v of headroom above v sw . as v s is reduced to .9v, excessive v bst droop decreases transistor driver headroom, as shown in figure 5(b). the power transistor can no longer be fully saturated (as evidenced by the round v sw turn-off corners), resulting in high power dissipation. when bootstrapping from a low- voltage output or supply, checking the bootstrap voltage is a good precaution. since the inductor current charges c 3 , the bootstrap circuit requires some minimum load current to function. figures 6(a) and 6(b) show the minimum input voltage required to saturate the power transistor and to produce a regulated output as a function of the load current. once started, the bootstrap circuit is able to sustain itself down to zero load. minimum input voltage 5.0 5.5 6.0 6.5 7.0 1 10 100 1000 load current (ma) input voltage (v) v out = 5v to star t dropout minimum input voltage 3.0 3.5 4.0 4.5 5.0 5.5 1 10 100 1000 load current (ma) input voltage (v) v out = 3.3v to star t dropout the minimum input voltage required to start and to operate before dropout (a) v out =5v (b) v out = 3.3v figure 6. figure 5. switching waveforms of a 10v to 5v regulator (a) sufcient bootstrap voltage drives the power transistor into saturation, minimizing power loss. (b) excessive droop in bootstrap capacitor voltage fails to keep the power transistor saturated near the end of its conduction cycles, causing jitters and low efciency. v sw v bst v in all traces 2v/div (a) 400ns/div v bst v in v sw 400ns/div all traces 2v/div (b) (a) (b)
4 SC4530 applications information (continued) feed-forward compensation a feed-forward capacitor c 4 (connected across the upper feedback resistor r ) is needed for stability. an initial estimate of c 4 can be found using equation ( 0) below: (0) the value of c 4 can be optimized empirically by observing the inductor current and the output voltage during load transient. starting with the initial estimate, c 4 is tuned until there is no excessive ringing or overshoot in the inductor current or the output voltage during load transient. mode transition and the fb pin if the upper feedback resistor r (figure 3, page 9) is large and is about the same magnitude as r 2 , then fast switching transients may couple into the fb pin, disturbing or delaying the transition from light-load operating mode to continuous-conduction mode (ccm). as described previously, the output ripple voltage is very low in continuous-conduction mode. delayed ccm transition extends the load range in which the converter produces larger output voltage ripples. this disturbance becomes more pronounced when v in is increased above 2 v and sw_out_il_28v to 3.3v@145ma_c4=22pf_onset ccm=150ma sw_out_il_28v to 3.3v@118ma_c4=22pf_c5=47pf_onset ccm=121ma i l1 200ma/ div v sw 20v/div 10 m s/div c 4 = 10pf, c 5 = 33pf v out = 5v, i out = 110ma ccm onset i out = 126ma (b) figure 7. switching waveforms of a 28v to 5v converter just before it enters continuous- conduction mode figure 8. switching waveforms of a 28v to 3.3v converter just before it enters continuous- conduction mode v out 50mv/div ac coupled i l1 200ma/ div v sw 20v/div 4 m s/div c 4 = 22pf, c 5 not placed v out = 3.3v, i out = 145ma ccm onset i out = 150ma (a) 10 m s/div v out 50mv/div ac coupled i l1 200ma/ div v sw 20v/div c 4 = 22pf, c 5 = 47pf v out = 3.3v, i out = 118ma ccm onset i out = 121ma (b) 1 6 4 r 10 8 . 6 c u sw_out_il_28v to 5v@174ma_c4=10pf_onset ccm=176ma v out 50mv/div ac coupled i l1 200ma/ div v sw 20v/div 4 m s/div c 4 = 10pf, c 5 not placed v out = 5v, i out = 174ma ccm onset i out = 176ma (a) sw_out_il_28v to 5v@110ma_c4=10pf_c5=33pf_onset ccm=126ma v out 50mv/div ac coupled
5 SC4530 when large feedback resistors are used. the regulator becomes insensitive to switching disturbances after it enters continuous-conduction mode. the operating mode transition can be significantly smoothed by filtering the fb node. a capacitor between fb pin and ground (capacitor c 5 , as shown in figures 3(a), page 8) serves this purpose. it should be chosen so that it improves mode transition without significantly slowing down load transients. switching waveforms of a 5v output regulator (figure 3(a), page 8) immediately before it enters continuous-conduction mode are shown in figure 7. the inductor current waveform appears to be more jagged without filtering. moreover, transition to ccm occurs at an output current of 76ma, instead of 26ma with fb filtering. figure 8 compares the corresponding switching waveforms of an output 3.3v (figure 4(a), page 9) regulator. applications information (continued) if the converter output voltage is .8v or less, or if r 2 is reduced to below 2k w , then c 5 will not be necessary. c 5 is also optional in figures 3(a) and 4(a) if the maximum v in never exceeds 2 v. bench testing shows that removing c 5 from these converters still results in acceptable transitional behavior, provided that v in < 2 v. c 5 can be estimated using the following empirical equation: () fb filtering has no significant impact on the output ripple voltage. however, it improves the converter efficiency by 0.25% to 0.5% around the mode transition point (figure 9). regulator efficiencies are slightly lower (< 0.25%) at light loads when filtering the fb voltage. positive values in figure 9 imply that fb filtering improves efficiency compared to no filtering. effect of fb filtering on efficiencies vs load current -1.0 -0.5 0.0 0.5 1.0 1 10 100 1000 load current (ma) efficiency difference (%) v in = 28v v out = 5v v out = 3.3v 4 2 1 6 5 c r r 10 8 c ? u efectfofffbffilteringfonfconverterfefciencyf(v in f=f28v) plottedfefciencyf=fthefefciencyfoffaffb-filteredfconverterf-f thefefciencyfoffthefsamefconverterfwithoutffb-filtering figure 9.
6 SC4530 reverse input protection consider a circuit board where the input power source supplies several dc-dc converters, including an SC4530 regulator with a large output capacitor. during power- off, the SC4530 regulator output may be held high by its output capacitor, while v in is discharged rapidly by other dc-dc converters. if v in falls to two diode voltages below v out , then the parasitic junction diodes inside the SC4530 (see figure 2, page 5) will draw current from the output through the sw pin to the input. if the load is light and the output capacitor is large, then high reverse current will flow, or even damage the internal circuits. figure 0 shows two protection schemes. in figure 0(a), a schottky diode d 4 placed at the input blocks the reverse current. this method has the disadvantage that it lowers the converter efficiency. a pn junction diode placed from the converter output to the input [(as shown in figure 0(b)] shunts the reverse current away from the part, thus protecting the part. this scheme is not suitable in a power supply system where a backup battery is diode or-ed with the SC4530 regulator output and with the SC4530 input grounded. board layout considerations in a step-down switching regulator, the input bypass capacitor, the main power switch and the freewheeling power diode carry pulse current with high di/dt (figure ). to minimize jittering, the size of the loop formed by these components must be minimized. since the main power switch and the freewheeling diode are already integrated inside the part, connecting the input bypass capacitor close to the ground pin minimizes size of the switched current loop. applications information (continued) out v l z heavy lines show the fast switching current paths in a step-down converter. the input capacitor should be placed close to the part for improved switching performance. figure 11. out v l z heavy lines show the fast switching current paths in a step-down converter. the input capacitor should be placed close to the part for improved switching performance. figure 11. SC4530 sw bst in bias out v in en gnd fb off on d4 (a) reversed input protection schemes (a) d 4 blocks the reverse current (b) d 4 shunts the reverse current from the part during power-of. figure 10. (b) SC4530 sw bst in bias out v in en gnd fb off on d4 1n4148
7 SC4530 shortening the traces at the sw and bst nodes reduces the parasitic trace inductance at these nodes. this not only reduces emi, but also decreases switching voltage spikes at these nodes. shielding the fb trace from the sw and the bst nodes with ground traces is a good precaution in mitigating switching transient disturbance. figure 2 shows an example of external component placement around the SC4530. the exposed pad should be soldered to a large power ground plane as the ground copper acts as a heat sink for the device. applications information (continued) u1 vout l1 vin c3 r4 r1 c4 r5 c2 r2 c5 c1 gnd sw figure 12. suggested pcb layout for the SC4530
8 SC4530 typical application circuits l1: coilcraft lps4018 c2: murata grm32er71h475k c1: murata grm31cr60j226k SC4530 sw bst in bias out 5v/0.3a r1 619k v in 6v - 30v 200k r2 l1 33 m h en c3 0.22 m f c2 4.7 m f gnd fb c1 22 m f c4 10pf off on c5 33pf c4=22pf c5=68pf verified 1/24/2011 c4=10pf c5=33pf verified 1/12/2012 sw_out_il_12v to 5v@300ma_c4=10pf_c5=33pf sw_out_il_12v to 5v@10ua_c4=10pf_c5=33pf sw_out_il_12v to 5v@60ma_c4=10pf_c5=33pf out_il_12v to 5v_ld_tran_0 - 300ma_c4=10pf_c5=33pf (a) 5v output step-down converter (b) switching waveforms of the figure 13(a) circuit. v in = 12v, i out = 10 m a (c) v in = 12v, i out = 60ma (d) v in = 12v, i out = 300ma (e) load transient. v in = 12v, i out is switched between 0 and 300ma figure 13. v out 50mv/div ac coupled i l1 200ma/ div v sw 10v/div v out 200mv/div ac coupled i l1 200ma/ div 40 m s/div (b) (c) (d) (e) (a) v out 10mv/div ac coupled i l1 200ma/ div v sw 10v/div 1 m s/div v out 20mv/div ac coupled i l1 200ma/ div v sw 10v/div 1ms/div 4 m s/div
9 SC4530 typical application circuits (continued) out_il_24v to 3.3v_ld_tran_0 - 300ma_c4=22pf_c5=47pf sw_out_il_24v to 3.3v@100ua_c4=22pf_c5=47pf sw_out_il_24v to 3.3v@60ma_c4=22pf_c5=47pf sw_out_il_24v to 3.3v@300ma_c4=22pf_c5=47pf (a) 3.3v output step-down converter (b) switching waveforms of the figure 14(a) circuit. v in = 24v, i out = 100 m a (c) v in = 24v, i out = 60ma (d) v in = 24v, i out = 300ma (e) load transient. v in = 24v, i out is switched between 0 and 300ma figure 14. v out 10mv/div ac coupled i l1 200ma/ div v sw 20v/div 4 m s/div (c) 1 m s/div (d) v out 200mv/div ac coupled i l1 200ma/ div 40 m s/div (e) SC4530 sw bst in bias out 3.3v/0.3a r1 332k v in 4v - 30v 200k r2 l1 33 h en c3 0.22 f c2 4.7 f gnd fb c1 22 f c4 22pf off on c5 47pf l1: coilcraft lps4018 c2: murata grm32er71h475k c1: murata grm31cr60j226k c4=10pf c5=47pf verified 1/13/2012 c4=22pf c5=47pf verified 1/26/2011 c4=22pf c5=47pf to reduce start - up overshoot 3/14/2012 (a) 400 m s/div (b) v out 50mv/div ac coupled i l1 200ma/ div v sw 20v/div v out 20mv/div ac coupled i l1 200ma/ div v sw 20v/div
20 SC4530 typical application circuits (continued) SC4530 sw bst in bias out 9v/0.3a r1 931k v in 10v - 30v 147k r2 l1 47 m h en c3 0.22 m f c2 4.7 m f gnd fb c1 10 m f c4 6.8pf off on c5 68pf l1: coilcraft lps4018 c1: murata grm31cr61c106k c2: murata grm32er71h475k SC4530 sw bst in bias out 2.5v/0.3a r1 205k v in 3.2v - 30v 200k r2 l1 33 h en c3 0.47 f c2 10 f gnd fb c1 47 f c4 33pf off on c5 47pf l1: coilcraft lps4018 c2: murata grm32er61h106k c1: murata grm21br60g476m c4=33pf c5=100pf verified 1/27/2011 c4=33pf c5=47pf to reduce start - up overshoot 3/14/2012 l1: coilcraft lps4018 c1: murata grm31cr60g107m SC4530 sw bst in bias out 1.8v/0.3a r1 93.1k v in 3.1v - 20v 200k r2 l1 33 h en c3 0.22 f c2 4.7 f gnd fb c1 100 f c4 33pf off on l1 changed from 22uh to 33uh 1/17/2012 c2: murata grm32er71h475k l1: coilcraft lps4018 c1: sanyo poscap 4tpe150maub SC4530 sw bst in bias out 1.23v/0.3a v in 3.3v - 20v l1 33 h en c3 0.22 f c2 4.7 f gnd fb c1 150 f off on c2: murata grm32er71h475k verified 2/17/2011 SC4530 sw bst in bias out 12v/0.3a r1 866k v in 14v - 30v 100k r2 l1 68 h en c3 0.22 f c2 4.7 f gnd fb c1 4.7 f c4 10pf off on c5 100pf l1 changed from 47uh to 68uh 1/16/2012 l1: coilcraft lps6225 c1: murata grm31cr61c475k c2: murata grm32er71h475k * lps4018 68uh saturation current is marginally low change c4 from 22pf to 10pf and c5 from 470pf to 100pf, 3/15/2012 figure 15. 1.23v output step-down converter figure 16. 1.8v output step-down converter figure 17. 2.5v output step-down converter figure 19. 12v output step-down converter figure 18. 9v output step-down converter
2 outline drawing C mlpd-w-8 3x2 SC4530
22 land pattern C mlpd-w-8 3x2 SC4530
23 SC4530 semtech corporation power management products division 200 flynn road, camarillo, ca 930 2 phone: (805) 498-2 fax: (805) 498-3804 www.semtech.com contact information ? semtech 20 2 all rights reserved. reproduction in whole or in part is prohibited without the prior written consent of the copyright owner. the information presented in this document does not form part of any quotation or contract, is believed to be accurate and reliable and may be changed without notice. no liability will be accepted by the publisher for any conse - quence of its use. publication thereof does not convey nor imply any license under patent or other industrial or intellec - tual property rights. semtech assumes no responsibility or liability whatsoever for any failure or unexpected operation resulting from misuse, neglect improper installation, repair or improper handling or unusual physical or electrical stress including, but not limited to, exposure to parameters beyond the specified maximum ratings or operation outside the specified range. semtech products are not designed, intended, authorized or warranted to be suitable for use in life- support applications, devices or systems or other critical applications. inclusion of semtech products in such applications is understood to be undertaken solely at the customers own risk. should a customer purchase or use semtech products for any such unauthorized application, the customer shall indemnify and hold sem - tech and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs damages and attorney fees which could arise. notice: all referenced brands, product names, service names and trademarks are the property of their respective own - ers.

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